I. Field of the Invention
The invention relates generally to communications. In addition, the invention relates to wireless telecommunications including cordless telephones.
II. Description of the Related Art
The cordless telephone has become a popular consumer good. The cordless telephone allows a user to untether himself from a wired connection to his local telephone line. Typically, a cordless telephone is comprised of two units: a base unit and a handset both of which are transceivers. The base unit connects to the public switched telephone network typically using a standard RJ-11 connector. The base unit provides a wireless connection to a handset. The handset is capable of receiving and transmitting signals over a wireless link to the base unit. The use of the wireless link allows the handset to communicate with the base unit.
Many cordless telephones operate as a time division duplex (TDD) system. In time division duplex, the base unit and the handset alternately transmit such that the units do not transmit at the same time. In a time division duplex system, the same frequency band can be used for both transmission and reception. By using time division duplex, the transmit and receive circuitry within each unit can share common components. In addition, each unit requires less internal isolation between the transmit and receive circuitry. For these reasons, a cordless telephone which operates using time division duplex can be cheaper, more reliable and yet produce higher quality audio signals than a full duplex unit. Even though the wireless link operates using time division duplex, audio compression techniques are used to provide concurrent bi-directional audio communication to the user. Therefore, even though the wireless link signals are time division duplex, the end user perceives simultaneous bi-directional audio communication.
In addition, cordless telephones typically use direct sequence spread spectrum (DSSS) modulation in conjunction with TDD. Spread spectrum signals used for the transmission of digital information are distinguished by the characteristic that their bandwidth is much greater than their information rate in bits per second. The large redundancy introduced by spread spectrum operation can be used to compensate for severe levels of interference. In addition, spread spectrum can be used to introduce pseudo-randomness into the signal. Transmission signals spread with a pseudo-random code appear to be random noise and are difficult to demodulate by receivers other than the intended receiver. In this way, a system which uses direct sequence spread spectrum is less vulnerable to accidental or deliberate reception by a third party. In this way, direct sequence spread spectrum, in conjunction with a scrambling scheme, provides a significant element of privacy in the communications channel between a handset and a base unit.
In a direct sequence spread spectrum system, data bits are modulated with a spreading sequence before transmission. Each bit of information is modulated with a series of chips from the spreading sequence. The number of chips per bit defines the processing gain. A greater number of chips per bit creates a greater immunity to noise and other interference. For example, in one common cordless telephone spreading technique, each information bit is modulated with a 12 bit spreading code. Because a cordless telephone using direct sequence spread spectrum has an enhanced immunity to noise and other interference, the cordless telephone handset may transmit a very low output power.
In a typical system, the spreading code might contain an even number of one's and zero's. In this way, the energy of the spread spectrum signal is minimized at and close to 0 Hz. For this reason, a baseband spread signal may be subjected to highpass or bandpass filtering with little effect on the information content. In a system in which each information bit is modulated with a 12 bit spreading code, a preferred spreading code can be chosen by examination of the spectral content of each possible 12 bit sequence which is comprised of six 0's and six 1's.
Prior to application of the spreading code to the information bit stream, the information bits may undergo a series of digital operations which further increase the performance of the system. For example, the information bits may undergo differential encoding in order to be more intolerant to an incorrect phase lock in the receiving unit phase locked loop (PLL). The information bits may be scrambled using a long scrambling sequence in order to further decrease the vulnerability of the system to interception.
Conventional cordless telephones utilizing direct sequence spread spectrum coding also use binary phase shift keying (BPSK). In a phase shift keyed system, information is carried in the phase of the signal. For example, in FIG. 1A, the binary sequence 1 0 1 1 0 is represented as a series of positive and negative voltage levels. In FIG. 1B, the same sequence has been phase shift keyed modulated. In FIG. 1B, two different phases are used to denote the two different digital values. Note that whenever the sequence transitions from a “1” to a “0” or from a “0” to a “1”, the phase of the signal in FIG. 1B transitions. Such a system is referred to as a BPSK system.
FIG. 2 is a block diagram showing a prior art BPSK architecture. This architecture may be used by both the base unit and handset. A digital mixer 21 (contained within the digital architecture) receives the digital data produced by a digital portion of the architecture which is not shown in FIG. 2. The spreading code generator 22 supplies the spreading code to the other input of the mixer 21. The digital spread spectrum waveform output from the mixer 21 is converted to an analog signal by a one bit digital-to-analog converter (DAC) 62. The analog baseband signal is then amplified by a baseband amplifier 60. After amplification, the signal is passed through bandpass filter 58. The bandpass filter 58 is employed to remove higher order harmonics contained within the baseband spread spectrum signal in order to avoid transmitting out of band energy. In addition, the bandpass filter 58 attenuates signal energy at frequencies at or near 0 Hz. Attenuation of the low frequency components of the baseband signal aids in suppression of the radio frequency (RF) carrier frequency component of the radio output. In another embodiment of the system in FIG. 2 the bandpass filter 58 can be replaced with a lowpass filter.
The filtered output of the bandpass filter 58 is modulated with an RF carrier by a mixer 56. The RF carrier is generated by a phase lock loop comprised of a voltage control oscillator (VCO) 44, a lowpass filter 46 and a frequency mixer/phase detector 48. During operation, the mixer/phase detector 48 is programmed by the digital architecture to control the VCO 44 to generate an RF sinusoidal signal at the selected wireless link center frequency. The signal produced by the VCO 44 is applied to the mixer 56 such that the output of the mixer 56 is a BPSK waveform at the desired RF transmit frequency.
The RF BPSK waveform is amplified by an amplifier 54. In addition, the BPSK waveform is amplified by a variable gain power amplifier 50. The gain of the power amplifier 50 is set based upon a transmit power level indication received from the digital architecture and converted to usable form by a power amplifier level control unit 52. The gain of the power amplifier 50 at the transmitter may be decreased as the path loss between the handset and base unit is decreased in order to conserve power. During a transmission period of the time division duplex operation, an RF switch 22 connects the output of the power amplifier 50 to a radio frequency lowpass filter 20. The output of the lowpass filter 20 is transmitted to the receiving unit over an antenna.
During a reception period of the time division duplex operation, a receive signal passes through the lowpass filter 20. The radio frequency switch 22 connects the output of the lowpass filter 20 to an RF bandpass filter 24. The output of the bandpass filter 24 is passed to a variable gain low noise amplifier 26. The gain of the low noise amplifier 26 is selected by an LNA gain level indication generated by the digital architecture. The gain of the low noise amplifier is decreased as the path loss between the base unit and the handset decreases in order to avoid saturation of the receive circuitry. In order to discern the phase of the received signal at the baseband, the received RF signal is down converted using an in-phase and quadrature component of the RF signal produced by the phase lock loop. The RF signal produced by the phase lock loop is shifted by 90 degrees by a phase shifter 42 before use in the quadrature receive path. The in-phase and quadrature components are applied to the mixers 28A and 28B respectively. The output of the mixers 28A and 28B are passed to bandpass filters 30A and 30B, respectively. The output of bandpass filters 30A and 30B are passed to variable gain amplifiers 32A and 32B respectively. The gain of the variable gain amplifiers 32A and 32B is set by a baseband gain level indication received from the digital architecture to control the signal level supplied to subsequent components. The output of the variable gain amplifiers 32A and 32B is converted to a digital representation by analog-to-digital converters (ADCs) 34A and 34B.
The output of ADCs 34A and 34B is sent to matched filters 38A and 38B via a phase rotator 36. The phase rotator 36 attempts to compensate for any frequency offsets affecting the received baseband signal. Although both the transmitting and receiving units have a PLL, the carrier signal produced by the receiving unit is never exactly the same as the carrier signal produced by the transmitting unit due to injected noise, reference frequency variations and other sources of errors. Any difference between the transmitter and receiver carrier signals modulates the resulting baseband signal produced by the receiving unit. The phase rotator 36 attempts to detect and correct for errors due to frequency and phase offsets which modulate the baseband signal.
The matched filters 38A and 38B perform the despreading functions. The despreading function removes the direct sequence spread spectrum modulation from the received signal. The outputs of the matched filters 36A and 36B is input into a BPSK demodulator 40. The BPSK demodulator 40 uses the amplitude of the output of each matched filter 38A and 38B in order to recover the transmitted information bits from the received signal. A differential decoding stage may also be used if the information bits have been differentially encoded at the transmitter.
Cordless telephones employing direct sequence spread spectrum modulation and time division duplex typically provide a usable data rate of 100 kilobits per second in a full duplex communication link. The full duplex communication link provides for high quality voice communication. However, such a system has many limitations which make it unacceptable for data transmission. For example, the DSSS architecture makes it very difficult to increase the usable data rate due to restrictions in the amount of signal bandwidth available in the 902 MHz-928 MHz ISM (Industrial, Scientific and Medical) frequency band utilized by cordless telephones under FCC regulations. In addition, typical time division duplex schemes employed with cordless telephones allocate fixed, equal time intervals for transmitting and receiving for the handset and base unit. Such an inflexible approach is inefficient for data transmission. Therefore, current cordless telephone systems have many drawbacks for data communications.